Selective sideband transmission and reception system



Sgpt. 16, 1952 o. E. NORGAARD 2,611,036

SELECTIVE SIDEBAND TRANSMISSION AND RECEPTION SYSTEM Filed Nov. 12. 1947 4 Sheets-Sheet 2 SOURCE E g 1 H. EAMPL IF/EH AND MIXER l. (MAL OSC/LL A 7' 0/? Inventor: Donald E. Norgaar'd, b 7mom His Attorney Sept. 16, 1952 SELECTIVE SIDEBAND TRANSMISSION AND RECEPTION SYSTEM Filed NOV. 12, 1947 D. E. NORGAARD 4 Sheets-Sheet 5 Donald E. Norgaard,

m a W His Attorney P 6, 1952 D. E. NORGAARD 2,611,036

SELECTIVE SIDEBAND TRANSMISSION AND RECEPTION SYSTEM Inventor: Donald E. Nor'gaar'd,

His Attorney.

Patented Sept. 16, 1952 2,611,036 s V. SELECTIVE SIDEBAND TRANSMISSION 'AND' 1 Y e e RECEPTION SYSTEM c,.Donald Er 'Norgaard, S cotia, N. Y., assignor to General Electric Compan a corporation of.

New York Application November-12, 1947,-- Serial No. 785,259

My invention relates to communication systems and, moreparticularly, to aisystem of simultaneous single side-band modulation of a radio frequency carrier wave by two sources of intellii gence and reception of the resulting composite signalv with apparatus which separates the two intelligence sources and supplies them to different output circuits. 1 n V 1 It is a primary object of my. invention toprovide a new and improved system for transmitting and receiving intelligence on two sidebands associated with a singlecarrier wave. g

It is known that single sideband signals ma be generated by combining two separate modulations of, a carrier signal, one, modulation: being carried out on a signaLof referencephaseand the other ,on a carrier signal-of thesame frequency buthavinga phase displacement of 90 from the first signal. In anapplication for United States Letters Patent, Serial No. 662,665, filed by Robert,

B.,Dome on April 17, 1946, and' assigned'to the 9 Glaiins (01. 179-15) assignee of this present invention, there is dis closed a system employing two electricalnetworks, each having an essentially-logarithmic phase vs. frequency characteristic of transmis sion, which system permits the translation of -a complex wave into two corresponding complex waves whose individual components ca n be made to have an approximately fixedphase relatione ship over any desired band of frequencies. Such a system is particularly well adapted for use in r a single sideband transmission system to produce two audio frequency waves having components;

of the same ma gnitudes asthe components of an input audio frequency signal, but a nearlycone stant phase relationship of any desired value between the 'corresponding components in the two output waves. 1 l

It is likewise known that different intelligence;

by one s'ource to produce a single sideband on eitheruside of. the carrier waveo'r modulated by in both the transmitters and receivers special.

band pass filters intricate in design andexpensive in construction. It is a further object of my in vention to provide a system of this type which requires no bandpass-filters, butiuinstead em-v ploys relatively simple circuits in both, thetransmitting and receiving apparatus.

It is still another object of my invention-to .7 DIOVlde a new and improved communication 'system capable of multiplex transmission and re'ception of any two voice or entertainment channels.

and which requires no greatersband width than either channel alone when a conventional double a,

J carrying sidebands a sideband transmission and reception system is employed. y 4

It is still another object of my invention to provide a new and improved communication system for transmittingbinaural sound, in which system a conventional radio. receiver operates satisfactorily and in which a specially constructed receiver, separates two intelligencechannels employed in the transmitting; systenrtoproduce,

binaural' or biacousticaleffects. v v

It is still another object of my invention to pro videa new and; improved receiver for use for single sideband, reception; of a double sideband transmission signallingsystem in which an inter; we

fering transmission channel onone side or the other ,of.;a desired signal may be ,elimina ted from the output,

.rtlis still anotheri'obj'ect-of m message vid a new and improved radio receiver, circuit v capable offireceiving conventional double :side band. transmission-signals in which improved receptionis effectedcunder adverse conditions of iading and interference.

It iskalfurther object videta'znew and improved distortionless demodulation :system;;whi'ch permits the usenof, simple receiver. o -It i'sa still further object of myinventionto provide a new and improved transmitting sys-.

term in which aca'rrier wave may be modulated one'source toproduce a double sideband signal or- ;doubly single sideband 'mo'dulated by two sources, and in -any' of-which the carrier{wave' may be suppressed :o'r exalted to any desired degree:

another. object of I my invention to provide a new and improvedexaltedcarrier deof my, invention .to pro it is another obj ect of i my invention tour prov-ideiareceiver forsuch a system capable of utilizing the transmitted signal to'provide output signals' corresponding to the modulating signals applied to thecarrie'r' wave at the'transmitter. I

It is "still another object I of 7 my invention td providea new=:-and improved radioftransinitting systemfin which the principal portion of the radi-f ated energy is-- composed of' use'ful intelligence One of the features of my invention is the use of a transmission system which operates to permit simultaneous transmission of two different signals and to permit simultaneous output of one of the signals on a first pair of output terminals and of the other signal on a second pair of output terminals. In the transmitten'the respective signals are each passed through two electrical networks, which translate each of the modulating signals into two corresponding com plex waves whose individual components have an approximately fixed phase relationship over the entire band of frequencies of the respective signals. The two complex waves formed from the respective signals are used to modulate respectively a first carrier wave of reference phase and a second carrier wave of the same frequency as the first wave but having a fixed phase displacement from the first wave of the same magnitude as that of the components of the two complex waves used to modulate the.

respective carriers. The respective modulated carrier waves thus formed from each of the si nals are combined to form two sets of side bands located on opposite sides of the carrier wave and which may be radiated from any suitable antenna.

-In the receiving system of'my invention, the

received wave is heterodyned with a locally generated carrier wave to produce a so-called intermediate frequency signal. The intermediate frequency signal is demodulated in an exalted carrier detector which provides two output signals comprising components of the two different signals, corresponding components in the two output signals being displaced in phase by a fixed amount. These output signals, in turn, are translated through two electrical networks which may be identical with those employed in the transmitteranda differential amplifier or equivalent balancing circuit in order to separate the two different signals and supply them to different output terminals. 1 I When the signals thus translated are derive from different microphones, for example, exposed to the same sound source, binaural or biacoustical effects may be obtained by simultaneous reproduction of the two translated signals or selection of either one of the translated signals may be made to reproduce the sidebandh'aving the lesser amount of interference. Alternatively, the system maybe used for transmitting and receiving two totally different signals with selective reproduction of either of the signals, or both signals simultaneously through separate output devices. 1 The features of my invention which I believ to be novel are set forth with particularity in the appended claims. My invention itself, however. both "as to its organization and method of operation, together with further objects and advantages thereof, may best be understood byv reference to the following description taken in connection with the accompanying drawing, in

which Fig. 1 is a circuit illustrating certain fun-- d'amental principles of an exalted carrier detector'employed in the communication system of my invention; Fig. 2 is a modification ofthe detector circuit of Fig. 1; Fig.3 is'a. block diagram of my system for selective sideband transmission and reception; Fig. 4 is a circuit diagram of the phase shift network employed in. the system of Fig. 3; Fig. 5 is the diagram of a modulator circuit employed in the system of Fig. 3; Fig. 6 is the combined exalted carrier detector circuit employed in the receiver of the system of Fig. 3; and Figs. 7 and 8 are circuit diagrams of differential circuits suitable for use in the system of Fig. 3.

Fig. 9 is a block diagram showing a modification of the system of Fig. 3, and Figs. 10l5 are vector diagrams illustrating certain operational characteristics of the system of Fig. 9.

Referring to Fig. 1, I have there shown a basic schematic diagram of a detector or demodulator circuit which comprises a pair of diodes I, 2 having their anodes connected together and coupled through a capacitor 3 to a source A of signals, designated by the rectangle marked with the legend 4. -A second source B of signals, designated by the rectangle marked with the legend 5, is coupled by means of a capacitor 6 to the cathode of diode I. A pair of resistances I, 8 of equal value are connected, respectively, between the anode of diode I and a common point herein-after.referred to as ground and between the anode and cathode of diode 2. The cathode of diode 2 is likewise connected to ground through .a capacitance 9 of such value that it has alow reactance and serves as a by-pass by both sources A and B, but has .a high reactance to low frequency signals occupying the frequency range from 0 to 15,000 cycles per second, for example. A tuned circuit In comprising an inductance and capacitance connected in parallel and resonant at the frequency of the signals of source B is connected between the cathode of diode I and ground. The circuit ID has high impedance-to the frequency of signals supplied by the source B and low impedance to signalsof other frequencies including both the harmonics of the frequency of source B and unidirectional currents. Capacitor 6 is of such a value that it has low reactance to signals from source B. likewise, the internal impedance of' source B is low at frequencies in the range which includes the frequency of sources A and B. In some cases where the feedback of harmonics produced in the operation of the detector of Fig. 1 to preceding portions of the apparatus is very small, the capacitance of circuit I0 may be eliminated and desirable operation obtained by the use of only the inductance of this circuit.

When operating under condition such that source A applies a continuous signal to the anodes of diodes I, 2 and source B applies no signal to the cathode of diode I, the action of diodes I and 2 is such that capacitor 3 is charged negatively to a potential approximately equal to the positive peak value of the voltage of the wave ap plied by source A. Under these conditions, resistors 'I, 8, together with capacitor 3, allow each of the diodes I, 2 to act as peak rectifiers. Under such conditions, resistors I and 8 have unidirectional potential impressed on them equal ap-' proximately to the positive peak voltage of the wave of source A, together with the alternating potential of the Wave of source -A. One terminal of resistor I is grounded and the other is maintained-at a potential which is the algebraic sum diodes at the same time as the signal from source A and :the signals of the two sources are isochronous, that is, are of the samefrequency andarein phase, the cathode .of ;dio.de.':l;varies itsrpoten tial'in phase with its anode. s The veifective-result is to reduce ithe 'spotential applied .by: source A across diode I by an amou'nt equalto the potential. of source B; Thus, the direct current flowin throughresistor 1 is reduced by an amountlcorresponding to the peak voltage of the signal from source B; .Thedirect current flowing in resistor 8, however, is unaffected by the application of signal'from source B so that the potential ofcone ductor I'l changes from zero toa positive poten-'- tial equal-to the peak voltage ofthesignalappliedby source B. The same action is obtained if the connections of diodes I" and 2 are reversed. i

If source B is adjusted to-have the same frequency as source A, butis 180 out of phase with source -A,' anegative potential of the same magnitude is produced on conductor llJTh'e forc going descriptionof the action-of the circuit of Fig.1 holds't'rue as long-as the peak 'voltageofi source Bis materially smaller than that of source Under conditions when source 13 is-adjusted'to have a frequency differentfrorn that of-sou'rce A, the-potential of conductor ll varies at a rate equal to the difference between the frequencies of sources A and B, provided capacitors 3 and 9 have sufiiciently high reactances to'this'difler ence frequency. This condition is realized when the frequency difference is a-small percentage of the frequency of sourceA and whenthe poten tial ofsource B is materially-smaller than that of sourceA.

"'If'it is assumed frequency of r V L and'that source B contains two frequencies and I i-6P):

where w1=21r times the frequency of the carrier wave-of source A'and (oi-p) is 2,1'r tir'nesfthefrequency of the lower sideband produced bylfnock ulating a carrier wave of the frequency V 21r l with a signal of the frequency (the'rn'thod of obtaining such a signal i s explained hereinafter) and that the twoidentical frequencies are isochronous, the potential of conductor H then comprises two principal components'. One of these components is a un potential'component equal tothe peak voltage-pflthe component M or t t a that source A operates at aof source B and the other component is a voltage having a frequency equal to the difference fre-' having a carrier frequency I e e 1 v Substituting the y ues izrll and 1 from h m put of the detector, accordingly, may be The latter component has a peak voltage equal to-that of the component frequency v 1 1;; (ti-p) s eB- V. a;

The foregoing process may be analyzed math: tic .1.::a f llow lfs e B contains W componentfrequencies ,v

and

i 2 1':- the voltage of source Bis v E2=E cos w1t+EM cos (on-wt (1) where M may be called the modulation factor.

Equation 1 repersents a single sideband signal c1}. v 'I\.- e a and a modulating signalfrequency If sourceAy on the other hand, contains only. one component frequency. .11

in'ph'ase" with that 0f-source B, the vo1tage of sourceA,-E1',maybewritten:

r H Ei=KEcosi'w1ta If K is 'a=numerica1- constant always much greater than unity, the output on conductor l I F of the detector circuit of Fig. 1 may the expressed as twice the product of th-tW0 signals divided "by KEorq y 1 a:

tions arid- 1 Tait-1 E P fiWiEMi Pi-W eEM e ieh 1 1 EM cos ua-WM Since capacitor: 9 is of such a jvalue that .it has lowjreactance' to frequencies suchas that; from the source A} it has an'even lower rcactarice to 'higherf frequencies. "Thus; the components E cos 25:11? and 'EM cos (2w1 2 )'t of'Equation 3" are virtually shortecircuited by capacitor 9, while the co mponents E and EM cos pt, being respec;

eir, ig id ci eb l"a d1 by s cin nents, are not affected by capacitor 9. The outtten:

'- f Detector output=E+EM cos pt ,7 1 .'j= E' +M e Thisoutput corresponds'toia mod ate V V which may. be of the form (1'+M,'cos "Since Eq iefi ea' l -lhla l ei n ei.ais a=' ..sies amt 7. signal carrying the intelligence of the single tone of frequency the foregoing analysis'shows that such a single sideband signal may be demodulated substantially without distortion by the detector circuit of Fig. 1 provided K 1.

It is apparent from the foregoing that the circuit of Fig. 1 operates to demodulate the wave of source B. This demodulation takes place whether or not source B contains any component of frequency the only effect under such conditions being the omission from conductor II of the unidirectional component created by the presence of component frequency "2 2 of source B:

In the operation of the circuit of Fig. l, the potential of source A is much greater than that of source B, so that electron current flows through diodes I and 2'only near the positive peaks of signal applied by source A. When the source B is adjusted to have the same frequency and to be 90 out of phase with source A, current flows through diodes I and 2 during only a short portion-of each cycle of the wave from source A, conduction occurring only whenthe cathode of diode I is near zero potential. Thus, a component of the signal present in source B having the same frequency as the signal of source A does not affect the potential of the conductor II as long as that component in source B is 90f out of phase with that from source A. The potential of conductor II is affected, however, by :any change in the phase relationship betweenthe signalsof like frequency in the, two sources vA and B, the two 90 points being spe-,

ciflc cases where the detector output is zero., If, for example, source B contains a component having the same frequency as source A and the phase of this component in source B is altered from its 90 point with respect to the wave provided, by source A in such a manner that the cathode of diode I is positive with respect to ground at the instant source A causes diodes I and 2 to;conduct current, the unidirectional potential" on conductor 'II shifts from zero to a value corresponding to the product of the magnitude of that component of the wave supplied by source B and-the sine of the angular shift from the null or 90 point.

From-the foregoing, if source A applies a signal KE cos wit to the anodes of the diodes in the detector circuit of'Fi'g. fl and source B applies only as'ignal E sin (witin) .to the cathode of device I, ,""t resulting potential on conductor II is ."Esin (:MiirE sin n. (5)

when; is the phase displacement"ofjtliesource B mm the 90 phase relationship originallycon- The foregoing analysis also indicated the manner in which the circuit of Fig-1 operates to produce a signal having a frequency equal to the difference frequency between sources A and B when the sources" are not synchronous. This difference *frequency' wave 'is' sinusoidal' if the wave form" of "source B is sinusoidal and of-a 8 frequency different from that of source A. Such a combination of waves might be encountered, for example, if the circuit were used in telegraphy or Teletype systems or for purpose of speech" inversion. In such systems, the source A usually would not be of the same frequency as the source B. Likewise, in the foregoing, while the wave of the source A has been described as an unmodulated wave, this wave may have a low percentage of modulation, such as the ripple encountered in any normal filtering arrangement, without affecting adversely the operation 'of that circuit. Under any circumstances, however, the wave form of the difference frequency is the same as that of source B solong as the voltage of source B is materially less than that of source A.

In Fig. 2 I have shown a modification of the circuit of Fig. 1 in which two properly interconnected detector circuits of the type shown in Fig. 1 are employed to develop a control voltage ineone of the detector circuits automatically to adjust the frequency and phase of one of the sources so that a condition of isochronism is 3 maintained automatically at the other detector over a limited frequency range. Since th two detectorcircuits maybe identical, component elements of the second circuit corresponding to those of the first are indicated by reference numerals which are primed.

In order to shift the phase of the signals supplied by source A to detector I by 90 before applying it to diodes I and 2', I provide a 90 phase shift network I2 which is interposed between source A and the coupling capacitor 3, As is pointed out later, the network I2 operates to maintain a 90 phase relationship between the two signals applied from source A to the two detectors over only a small frequency range. Source B of Fig. 2 may be identical with the source B of Fig. 1. It is directly connected to the cathodes of detector diodes I, I so that iden tically the same signal is applied by source B to the two detectors.

I also provide means to adjust the frequency of source A over a narrow range in accordance with a control potential developed at the output terminals of the detector employing diodes I and 2. This means comprisesan electronic reactance circuit -or equivalent tuning device I3 connected between source A and an output terminal-of a low phase angle between the signal from source A and a corresponding component of source B synchronous with the signal present in source A. Thus, the unidirectional potential on conductor I6 connected between the low pass filter and tuning device I3 is the same as that of conductor II, there bein'gno possibility of unidirectional potential on thisconductor unless sources A and B have components that are synchronous. of source A is a function of'the potential of conductor I6 so that the loop circuit comprising source -'A, diodcs'l and 2, resistor I4, capacitor I5, and tuning device I3 may be made to operat in such a manner that the phase of the signal generated by source A is substantially displaced from that of a component of the signal from source B within the "range of frequencies throughout which tuningd'evice I 3 'controlsthe frequency of source A in accordance with the potential on I conductor I6. The deviation from 90 phase relationship between sources A -and B may be made very small by choosing'a value of E in Equation The frequency When expanded, this becomes:

9 and a control sensitivity for tuning device l3 such that the angle A of Equation 5 is essentially zero for a range of several kilocycles in the frequency of either source A or source B.

- The values of the low pass filter -elements l4, I5

are so chosen that this filter permits only gradual changes of potential on conductor I6 despite the instantaneous magnitude of voltage on conductor II. The voltage across capacitor I5 may be considered as the average of the voltage on conductor II, this average being taken over a period sufficiently long to mak the frequency of source A unresponsive to audio frequency signals which may appear on conductor II, but controllable nevertheless by the relatively long-time average of the potentialon conductor I l.

Source A may be considered as a locally generated carrier signal whose phase is definitely related to a component of the signal whichm-ay be present in the source B. An important, factorin the operation of a receiver employing the detector circuit of Fig 2 is the operation of the tuning device 7 H to maintain automatically a desired relationship between the'frequency and phase of source A'and that of a'selected component of the signal supplied by source B. When the phaseof the signal from source A appearing at its output terminal is displacedjby 90 from the phase ofa signal appearing at the output terminal of source B, then the signal at the output of the 90 phase shifter I2 is either in phase or 180 out of phase with the signal supplied by source B. Either phase relationship may be obtained simply by using the .phase shifter l2 either to advance or retard by 90 the phase of the signal supplied to diodes l and 2 by source A, dependin on the signal conditions desired at the anodes of diodes l, 2'. Of course, once a particular relationship is chosen, the phase relationship between the signals-appliedto the fourdiodes l, 2, I", 2' remains fixed as long as sourc A remains synchronized with source B through the action of tuning device i3. I Y i I A double sideband signalapplied to the oathodes of devices l' by the source B may be written:

Since, through the action of tuning device l3,, JsourceAis maintained at 90 phase relationship with respectto the carrier component B cos wt, the signal at the output terminal of source A is E; sin wt, where E is thepeak voltage of source A 'w-is-21r X carrier'frequency, and p is 21 X; modulation frequency." The signa1 applied-tothe anodes of diodes I312, through the action of th 9,0;phase shifter I2 is then E1 cos at 1 The detector outputonfconductor H is:

If only low frequency terms are considered, it is apparent that this output'has a value of zero.

In contrast, the output of the detector on conof source B, the signalsupplied to the 90 phase When expanded, this becomes:

apes mJF iI-H EOS as- 9 N 4 7 cos pt+cos(2w+p)t+cos pt] (10) If only low frequency terms are considered, the

detector output is: V

The term E in Equation 11 is the unidirec- 'tional' potential proportional to the carrier strength E and the term EM cos pt is the demodulated double sideband intelligence of Equation '6. The foregoing analysis shows that no signal exists on conductor H and that a signal proportional to an initial-signal of modulation (1+M cos pt) is present on conductor H.

'- "-When the outputfsignal supplied by source B to the cathodes of *devices- I, 'I' is a single sideband modulation for the lower sideband' associated with the carrier wave, the equation for such a signal may be:

When source A is synchronized with the component frequency shifter I 2 is:

- E1 sin (wt-45) i (13) due to the fact thatsource A lags thesignal from sourceB by Under such conditions, the signal impressed on the anodes of diodes I,

*2 is I E1 sin wt+45) (14) The resultant signal on conductor I I is:

zwfi sin (wt-t4?) +EM cos (wp)'t sin (wtl5)] (15) When this-expression is expanded and only low frequency terms considered, it becomes:

VEE+EM sin (pH-45) (18) vfico 90+EM sin (pt-45): I

' EM sin (pt-[45% (16) The signal on conductor H for this condition s: 2 WE sin (wt+45) +EM ,cos

When this expression is expanded and only low frequency terms considered, expression 17 becomes:

/2E+EM sin (pt+45) (18) It may be noted from the foregoing that the demodulated intelligence appearing on conductor 1 I has a 90 lagging phase relationship with that on conductor II. This follows from the fact that the potentials supplied respectively to the anodes of diodes l, 2, and the anodes'of diodes I, 2 were displaced in phase by 90. Concerning the action of the detector circuit of Fig. 2, it may be stated generally that in the demodulation of a single sideband signal by means of two circuits supplied with carrier signals having relative phase displacement, the demodulated output signals from the two circuits have the same relative phase displacement. i i

The detector circuit of Fig. 2 may be employed likewise to demodulate a signal modulated with Due to the operation of the tuning device l3, the signal in the output of source A is E1 cos t, where the value of E1 ismade much greater than a E. This signal is abritrarily chosen to lead the carrier signal E sin wt of Equation 19 by 90. In contrast, in Equation 13, the signal of source A was stated to lag that of source B by 90. Either mode of operation may be used but, once a mode is chosen, due account of the fact must be taken in subsequent analysis. 7

Accordingly, through the'action of 90 phase shifter I 2, the signal applied to the anodes of diodes I, 2' may be'Ei sin t. The demodulated output appearing on conductor H is:

2[E sin wt+M [cos (w-p) tl+ M lcos (w+q)tllcos wt (20) When only low frequency terms are considered,

this becomes:

=Mp cos pt+Mq cos qt (21) E+Mp sin pit-M sin qt (22) The unidirectional component, E, may be disregarded, since it is normally separated from the intelligence bearing signals p and 'q by means of coupling networks in a well-known manner.

From the foregoing, it is seen that the demodulated intelligence 10 on conductor ll leads that on conductor H by 90, while the demodulated intelligence q -on conductor ll lags that on conductor ll by 90.

In accordance with my invention, the signals on conductor H, when the detector is used for demodulation of "a carrier wave modulated by two such sources of intelligence p and q, may be supplied to one of a pair of networks of the type disclosed in the above-mentioned copending application, Serial No; 662,665, and the signals on conductor .l l may be supplied to the other network of the pair. The resultant signal at the output terminals of the first of such a pair 'of phase shift networks, which may be called an a network, is:

v where 11,) is the phase shift at the frequency and a is the phase shift at the frequency by the shift of the a network when operatingat frequencies 12 and The sum of the signals at the output terminals of the a and [3 networks is:

The difference of the signals at the outputs of the a and ,8 networks is:

From the foregoing, it is apparent that the two detector systems comprising the respective pairs of diodes 1, 2 and 1', 2 and their associated apparatus shown in Fig. 2 respond to a double'single sidebancl modulated carrier wave in such a manner that, when each supplies its output signal to a phase shift network system having the property of shifting phase in one portion thereof more than in the other over a wide band of audio frequencies, the simple algebraic sum of the two circuits contains one of the modulating signals and the difference contains the other modulating signal. While, in the foregoing discussion, only two frequencies,

have been considered, the symbols p and q can each represent any composite intelligence. Thus, intelligence p may be composed of any combination, such as 211, p2, pa, etc., where 121, p2, p3, etc. are the Fourier components of a complex Wave. Likewise, intelligence q may consist of Fourier components of a complex wave representing q. From the foregoing discussion, it is likewise evident that the demodulation circuit of Fig. 2 does not create intermodulatlon components, such as (pi-p2), (pip3). etc. or (pi-qr), (pH-qr), etc., none of which comprise the original intelligence of either modulating signal.

It will be shown from the description of the circuit of Fig. 3 to be given later that the intelligence derived by the circuit of Fig. 2 is that which is used in a transmitter for modulating the carrier wave to produce upper and lower sidebands. In systems where the sources A and B are not synchronous, it is of course necessary to remove the conductors I6 connected between the output circuit and the source A. Alternatively, if the connection comprising the conductor 16 and the tuning device I3 is made from the output terminal indicated to the source B and the source A comprises a crystal controlled oscillator, the circuit arrangement otherwise the same as Fig. 2 may be employed "to control the frequency of the source B.

In Fig. 3, I have shown the block diagram of a complete system for simultaneous transmission and reception of two channels of intelligence 12 and q on a single carrier wave. In the circuit of Fig. 3, the rectangular block numbered H and bearing the legend channel p mayrepresent a source of modulating intelligence which may occupy a band of frequencies from 30 to 10,000 cycles per second or greater, depending on the "prise one or more stages of amplification 'design'of the a and 3 networks employed. The

as channel 1) or a totally diiferent source. The

oscillator l9 bearing the legend to may comprise any suitable source, of carrier frequency,v such as.

a crystal-controlled oscillator. The signals of channel p are supplied to phase shifting networks 20, 21 labelled, respectively, a and 3, and which may be of the type described in the abovementioned copending application, Serial No. 662,665. The relationship between the networks 20, 2| is such that 5-a=90 for any modulating frequency between 30 and 10,000 cycles,where p is the phase shift of the network 2|, and a is the phase shift of the network 20. Similarcand 5 networks 22, 23 are connected to the output of the source labelled channel q. In order to select the sideband for the modulating signal q different from that selected for the modulating signal 11, the output leads from 23 to modulator 21 are reversed so that in effect a polarity reversal is obtained without a corresponding time delay.

This. is desirable in event-binaural signals are. to betransmitted inv separate sidebands in which -.case the time delays in passing from the. modulating sources I! and I8 through the respective phase shifting and modulator channelsto the power, amplifier are the same for corresponding frequency components. This insures that the relative phasingof the original binaural signals is retained for reproduction in a receivingsystem to be described shortly. The outputs of the networks 20, 2| are connected, respectively, to modulators 24 and 25, and the outputs of networks 22 and 23 are connected, respectively, to modulators26, 21. Modulators 24 and 26 are supplied with carrier waves or from the source l9,

while modulator 25, 21 are supplied with carrier waves a: which have been displaced in phase 'by 90 through the action of a phase shifter 28. The outputs of modulators 24-21, together withthe carrier wave a: amplified in an amplifier 29, are supplied to a power amplifier 30 which may comat'or around the carrier frequency of "The output ofthe power amplifier 30 may be supplied to a transmitting antenna 3l or any The receiver circuit of the system of Fig. 3

comprise an antenna 32 or any other suitable input device which couples a :portion. ot the energy from the output of power amplifier 30 to the input of a radio frequency amplifier tuned to a frequency of and mixer 33. In addition, there is supplied to the mixer portion of the device 33 a locally gen: erated heterodyning wave derived from an oscil-V lator 34. If the oscillations of oscillator 34 have a frequency differing from on a second pair of output terminals, provided by'a'n'amount' I 1I@ the output. of the amplifier and mixer 33 consists (among others) of modulated signals having an intermediate carrier frequency of Intermediate frequency signals-after a pliflcation in an amplifier 35, are supplied to;;an'exalted carrierdetector 36 which may comprise, for example, the circuit of Fig. 2. Since the intermediate frequency amplifier, 35 provides signal wave at the new carrierfrequency H glidl l i.

including upper and."lower s i debands separately bearing intelligence of-channels p and q,in employing ;the circuit arrangementof- Fig, 2" it b'ecomes necessary that the-source A showmin block 4 provides; waves-ofafrequency I Thus b1ock136 of Fig, '37mc1udes tiie maria arirangement, of ;Fig. 2 wherein source, B, shown'flin block 5 comprises'theioutputof the I513. amplifier 35, and source A shown in .blockl comprises a source of localio'slcillations' at=a frequency The output conductors of detector 35,; are numbered, respectively, I I, ll are connected to (land 5 networks 31,- 38 similar ,to thenetworks 20, 2| and 22, 23 employed in the; transmitter. The output currents of-the a andie networks, in turn, are combined in a differential amplifier 39. Preferably, the differential amplifier is of the type having two outputs, one of which comprises the algebraicsum of its' 'two input signals, and the other, the algebraic difference of its two input signals. The two outputs of the differential amplifier are connected, respectively,-to amplifiers 40, 4|. Amplifier 40 may be an audio amplifier for the channel p, and'a'mplifier' M,- for the channel q. At-the output terminals of the amplifiers 40," 4| appear, respectively, intelligence p and intelligenceq;

In the operation of the systemof Fig. 3, simultaneous transmission of intelligences p andq is effective to permit simultaneous output ofintelligence p on'one pair of output terminals and q the relationships of the various signals required by the Expressions 19 and 2326 are observed. In the transmitter of this system, these relationships are observed if the a ands networks maintain exactly phase relationship over anydesired range of modulation frequencies in transmitting the signal described by Expression 19 and if an exactly 90 phase relationship is maintained between the carrier signals obtained from the output of the oscillator l9 and thef-"90phase shifter28. It is "further required that the amplification and transmission be carried out by perfectly linear transducing elements between the modulation stages 20-21 and the demodulation stage 36.

are linear in their amplitude transfer characteristic. In the exalted carried detector 36 of the receiver, a fixed phaserelationship of 90 is desired for the locally generated oscillations of the source A with respect to the incoming carrier signal. To this end, the phase shifter I2 should provide an exactly 90 phase shift at all times. Finally, the circuits of the detectors comprising the diodes I, 2 and l, 2 preferably should have identical phase shift characteristics over the audio band of frequencies. I

While none of the foregoing desirable condi- "tionscan'be entirely satisfied in practice; I have found that the greatest difliculty is encountered in maintaining the exact 90 phase relationship of the a and 3' networks. While the performance of the receiver alone or the transmitter alone is governed largely by the characteristics of such a and B networks, I have found that it is possible to transmit an imperfect double single sideband signal and receive this signal on an imperfect receiver, provided certain specific-relationships are maintained. In other words, I have found that it is not'requiredthat, in the transmitting system, all the sideband energy of a band of frequencies be confined entirely to one side, or the other of the carrier frequency. I have likewise found that the receiver is not required" to be responsive only to sideband energy one side of the carrier wave, but may be responsive in a particular mannerto sideband energy which lies on the other side of the carrier wave. An explanation of the operation of the overall communication 1 system of Fig. 3, even when the a and 5 networks employed therein as not maintain an exact 90 phaserelationship 'overtheir entire range of intelligence frequency may best'be obtained from the following analysis. If the intelligence p comprises a sine wave having a frequency of anddntelligence q likewise comprises a sine wave having. a frequency of where both intelligences are confined to a frequency range from 30 to 10,000 cycles, they may be expressed as the following functions of time:

Intelligence p: Mt sin (ptap) Intelligence q: M 1; sin (qt-as where Ms is the amplitude of the wave 12; Mq is .the amplitude of the wave q; t. is the time. in. -.seconds;' up isthe. phase shift of the a network 'at theffrequency V and 11 is thephase shift of the a network at the frequency (ptdp+dp) =Mp sin pt thatof the a. network 22 is:

M, sin (qt-a +c -)'=M sm qt (so) 4.16 On the other hand, the; output of the ,8 network 21 is:

and a is defined-by the equation:

Inother words, a is the deviation from the ideal phase relationship assumed in former analysis.

Similarly, the output of network 23 is:

} M cos (qt-[tq) (31a) where q=flq-a'q90 (32a) in the manner of. Equation 32.

If modulator 24 is a balanced modulator sup plied with push-pull signals. of both carrier and modulating signal, the wave of its outputcurrent is described by the following: equation:

sin wm+M sin when wt(l-M sin pt) Similarly, since sin (ot+90)=coswt, the output current of modulator 25 is:

The equation of the output current of modulator 26 is:

Similarly, it can be shown that the output current of modulator 21 is:

The combined output of modulators 244! is the algebraic sum of Expressions 33 to 36. This combined output of the modulators is:

It is evident that, when 6 20 and a zo, Expression 37 represents two perfect single sideband signals having the same carrier frequency,

intelligence p being carried as the lower sideband and intelligence q being carried as the upper. In Expression 37, intelligence p is not confined wholly to the lower sideband nor is intelligence q confined wholly to the upper sideband, except when 61 :0 or 6 1:0, depending upon the signal considered.

The dependence of the ratio of the undesired to the desired sideband upon the error angle a may be shown by thefollowing considerations.

17 In the case of intelligence :0, the desired sideband is:

The ratios shown in Equations 40 and 41 indicate the performance of the transmitter alone, that is, the degree of perfection of the signals generated. The same ratios may be shown to hold true for the receiver alone and are a measure of the receiver performance or its ability to reject unwanted signals which occur in an unused sideband range.

The amplifier 3B of the transmitter system of Fig. 3 combines the signal output of the four modulators 24- 21 with a carrier wave of desired strength supplied by amplifier 29. The complete composite signal supplied to the amplifier 30 is,

therefore:

where E is the amplitude of the carrier signal, and the other terms are as defined previously.

In the receiver, thecomplex signal of Equation 42 is converted in the amplifier mixer 33 by heterodyning with oscillations generated locally in oscillator 34 to produce an intermediate frequency wave. The equation of the intermediate frequency wave is the same as Equation 42 with the exception that 401 is substituted for u, where ml is the intermediate frequency.

The circuit'of the exalted carrier detector 36 is basically the same as that shown in Fig. 2. The source A of that detector comprises a second locally generated oscillation which is made synchronous with the carriercomponent of the incoming signal. From the previous explanation of the operation of the circuit of Fig. 2, it is evident that in the exalted carrier detector 36 the phase of the output signal of source A is displaced by 90 from that of the carrier signal E sin wit supplied along with all other terms indicated in Equation 42 to the cathodes of diodes I, l,wi

being the intermediate frequency. The output signal of source A, therefore, may be written:

KEc 005 out (43) where K l and A of Equation is zero. The demodulation achieved by the diodes l, 2 in the 18 exalted carrier. detector 36 provides an output signal on conductor H whose expression is:

If only low frequency terms are considered, this becomes:

Equation 45 shows the simple algebraic difference of signals proportional to the outputs of networks 2| and 23 of the transmitter. Each of the two components of the signal is proportional to the output of a respective one of the ,6 networks in the transmitter and appears on the output conductor H of the detector 36 of the receiver. In similar manner, demodulation by diodes l', 2' of the detector 36 may be shown to produce the low frequency components on conductor H whose expression is:

This output is the simple algebraic sum of the signals produced by the networks 2|], 22 of the transmitter, namely, the algebraic sum of Equations 29 and 30. In the receiver the signals of Expressions 45 and 46 are supplied, respectively, to a and 5 networks 31, 38, which networks are identical with those employed in the transmitter.

The signal applied to the a network 31 is subjected to a phase shift of up for a frequency of and a phase shift or aq for a frequency of When the signal of Equation 45 is applied to such an a network, the output of the a. network 31 is: r

A signal applied to a 13 network is subjected to a phase shift of fip for a frequency Accordingly, when the signal of Equation 46 is applied tothe 5 network 38, the output of the network is:

. '19 In this expression, the unidirectional term is neglected, since thisterm preferably is removed by filtering before Ithe'a'signal is applied to the 5 network 38. When the values from Equations 32 and 32a for Li and fi in terms of up and (Lu, respectively, are substituted in Equation 48, this equation becomes: I j

In the receiver, the signals from the a and 8 networks 31 and 38 are supplied to a differential amplifier 39 which, in turn, provides to amplifier 40 the sum of the two signals represented by Equations?! and'49and to amplifier 4|, the difference of these'two signals.- Thus, the signal supplied to amplifier 40 is:

Similarly, the signal supplied to amplifier 4! may be shown to be:

These Expressions 50 and 51 show that, despite the imperfections inthe a and )3 networks, the imperfect transmission received on the receiver cos employing the-imperfect a. and 8 networks may ZMp sin (pt-ta +2Mq sin (qt+ ,q) (52) where the unidirectional term is disregarded, and the output of the network is:

substituting the values of Bp and fi in terms of up and do. Since cos (A+90 ):-sin A, Equation 53 may be written:

When the complex currents of Equations .52 and 54 are added in the differential amplifier 39, the output at one terminal of the amplifier is:

Likewise, if these currents are subtracted, the output at the other terminal of the amplifier is:

used in the transmitter or receiver.

From the last two equations, it is seen that, if a and 6 are not zero or if the connections to the a and p networks .31, 38 are not as shown in Fig. 3, neither signal is obtained separately from the other. The ratio of the undesired signal at respective outputs of the differential amplifier to the desired signal in such a case is, of course, proportional to the strength of the undesired signal, inversely proportional to the strength of the desired signal, proportional to 2 sin 6 and inversely proportional to 2 cos 6q,p.

Referring .now to the operation of the power amplifier 30 in the transmitter of Fig. 3, if this amplifier is considered to be a single stage high efficiency class AB1 or class ABz amplifier having one tube operating within the range to satisfy the requirements of this type of operation, as an approximation the transfer characteristic of such an amplifier may be written:

Output voltage=i ZL=K(e 2 (57 where ip=the instantaneous plate current,

ZL=the plate load impedance,

K =a constant for the tubeused, and

e =the instantaneous signal voltage applied.

The term e normally is composed of the sum of a bias potential C and signals to be amplified. If the signal to be amplified is one such as described by Equation 19, the signal applied to the grid'is:

- 6v=C+E sinwt-l-M lcos (ii-2 t] 1 --|-Mq[cos (w+q)tl The square of this expression is:

When a filter is used to remove all frequencies other than thosenear the'output of'such an amplifier becomes:

or areplicaofthe original signal.

From-the foregoing, it is apparent that an amplifier which permits linear amplification of a double sideband signal, also satisfactorily translates a double single sideband signal. This, of course, is true whether such an amplifier is One type of non-linearity, however, re-introduces the other, or unwanted, 'sideband. This non-linearity, which arises inamplitude limiting circuits, is equally undesirable for double sideband amplitude modulationsignals. 'It is apparent, there- 'fore, that in my improved communication system no greater restriction is placed upon the distortion permitted in an amplifier than is placed upon'the same amplifier when used for translat- "ing conventional amplitude modulated signals.

From the foregoing discussion, it is apparent that my improved communication system may be made to give theoretically perfect separation :of the channels of intelligence transmitted, even though the a and ,3 networks of the system do not maintain an exactly phase relationship and the amplification and transmission are not terference.

' carried out by perfectly linear impedance and transducing elements. On the other hand, it is well known that substantially constant 90 phase relationship may be maintained betweena selected component of thesignal B and the source A employed in my detector circuit. In normal operation, the carrier frequency of my transmitter is fixed and stable adjustment of the carrier phase may be provided. Accordingly, it is apparent that my communication system illustrated in Fig. 3 and described above is capable of operating satisfactorily to separate two different intelligence channels. I have found in practice that a separation between the two channels in the order of from 40 to 50 decibels may be maintained in this communication system.

One important application of the communication system of Fig. 3 arises when the channels 12 and q are microphones or other pickup devices which are exposed to the same sound field in a manner similar to the two ears of a human being. The resultant signal which is delivered to a radiating antenna 3| cannot be distinguished from a conventional double sideband signal by receivers of conventional design. A conventional receiver operates on such a signal as though the two microphones were mixed together and the resulting signal modulated in a carrier by conventional means. However, the improved receiving system of Fig. 3 is capable of resolving the information picked up by each microphone and supplying two different acoustical devices with corresponding signals. This is shown in Fig. 3 in which the respective channels 40 and H are illustrated as supplying their respective signals to diiierent loudspeakers. When such dual transmission and reception is employed, binaural reception is obtained when headphones are used in place of the speakers. Similarly, biacoustic or stereophonic reproduction is obtained when loudspeakers are employed. In such a system, I have discovered that the binaural system imparts a degree of realism to any transmission which makes it difficult to distinguish sounds transmitted by the system from those of local origin and which do not pass through any artificial transmission means. Likewise, I have found that the biacoustic reproduction enhances the reproduction of sounds, the effect being largely dependent upon the acoustical conditions at the location of the reproducers. striking andgimparts to intelligence transmitted by such a system a realism never heretofore encountered by any artificial transmission means employing but one intelligence channel and which is obtained otherwise only when the listener is present in the locality of origin of the sounds or intelligence signals.

An important advantage of the. communication system of the type shown in Fig." 3 is that it may be used for broadcasting binaural sound, while still permitting the conventional receivers now in existence to operate in a normal fashion. At the same time, the special receiver described operates to provide binaural or biacoustic reproduction with the added improvement of realism in the transmitted intelligence.

Another important advantage of my improved communication system is that two channels of intelligence, such as the channels p and q, may be transmitted and received without mutual in- In such a closed system, moreover, it is not required to transmit an appreciable amount of carrier power, since the detector circuits of Fig. 2 functionto-provide substantial- The binaural effect is quite 1y distortionless operation, even with zero transmitted carrier. As a result, the principal portion of the radiatedenergy in-such a system is composed of useful intelligence bearing sidebands.

In a system of this type *which transmits zero carrier wave, a high order of frequency stability is required in the receiver for the local oscillator 34 and the oscillation source A in the exalted carrier detector and in the transmitter for the carrier frequency. I have found, however, that, when the level of the carrier Wave transmitted is approximately 10% ofthat normally employed in an amplitude modulated transmitter, reliable operation of the receiving circuits is insured, especially for the frequency and phase controlling device ill of the detector circuit of Fig. 2. By the expression carrier wave normally employed is meant a carrier wave of such magnitude that the combined negative peaks of the modulating signals are never greater thanthe peak of the carrier. Thus, such a condition is obtained when a conventional carrier wave is modulated less than 100%, and a conventional receiver supposedly operates satisfactorily on such a signal. While the transmission of small amounts of carrier permits practically all the transmitted power to be used for intelligence carrying sidebands, receivers which are not provided with exalted. carrier detectors are not capable of demodulating such a signal without distortion. For example, when two channels, such as channels 11 and q, are transmitted in too great a proportion with respect to the carrier, a conventional receiver combines these signals to produce distortion. In other words, when a small amount of carrier wave is transmitted, a double single sideband receiver is required to prevent distortion or intermodulation of the p and q channels. In such a case, my improved receiver causes substantially no distortion, permitting maximum utilization of practically all transmitted power..

In Fig. 4, I have shown typical a and ,8 networks, such as the networks 2t23 employed in the transmitter of Fig. 3 and the networks 3'5, 38 employed in the receiver of that figure. The network of Fig. ,4, which'is one of those disclosed in the above-mentioned copending application, Serial No. 662,665, comprises an input terminal 42 to which is supplied audio frequency voltages from which are to be derived twooutput voltages characterizedby the fact that the phase difierence of the two output voltages at any frequency is maintained substantially constant at any desired angle, .being utilized in the circuit of Fig. 3, as the input signal is varied over a wide range of frequencies. The input voltagesare illustrated as connected across a potentiometer 43 through a capacitor 4 3. A variable tap 45 on potentiometer 43 is connected to the input electrode oflan electron discharge device 46. The cathode and anode of the device at are connected, respectively, to ground and a source of unidirectional potential indicated by the legend through resistors M, 38 having equal values of resistance. In the conventional phase inverter circuit thus far described, voltages of equal amplitude, but opposite polarity, are available with respect to ground at the anode and cathode, respectively, of the device 45.

A phase shifting network comprising capacitor 49 and resistor 50 is connected between the anode and cathode terminals of device 45 to provide a voltage shifted in phase with respect to the input terminal 42 which may bev supplied directly to the control electrode of asecond electron discharge device Device 5 ltlikewise has its anode and cathode .connected, respectively, to positive operating potential and ground through matched resistances 52, 53 and has a phase-shifting network comprising capacitor 5% and resistance '55 connected between these electrodes.

In a similar manner, a third electron discharge device 56 has :its anode connected to the positive operating potential through a resistance 5'! and its cathode connected to ground through its resistance .58. A third phaseeshifting network comprising capacitor '59 and resistance 60 is connected between the anode and cathode of device 56. At the output terminals of the circuits of each of the devices .46, 5|, and 56, a successive shift in-phase of the output potential with respect to the voltage at the input terminal 42 is obtained, the total number of such phase shifting stages depending upon the band width desired for the a or 3. network. The output voltage of the device 56 is connected to the control electrode of a phase inverting electron discharge device :61. connected to the source .01 positive potential through a resistor 62 and the cathode is .connected to ground through a resistor 63, the resistance values of resistors 62, .63 being equal. Output terminals 54, 65, connected respectively to the anode and cathode of the device 61, provide two complex voltages of opposite polarity. When the product ofthe capacitor and resistance forming the phase shifting circuit of each stage has a definite predetermined ratio to the product of the capacitance and resistance of the phase shifting circuit of the next succeeding stage, the phase shift with frequency of the component :wave varies approximately logarithmically.

While, in this phase shifting network, if it is desired to increase the bandwidth over which the network operates to maintain the above mentioned logarithmic phase characteristic, it is necessary only to add successive stages having components selected in the manner described, I have found that abandwidth adequate for high quality speech and music is obtained by the use of but three stages of the circuit shown in Fig. 4.

'I have further found that the'stages bearing the predetermined ratio of resistance-capacitance product may be arranged in any desired sequence and need not be successive stages as illustrated. In constructing an a or 5 network in accordance with this circuit, each coupling stage must be well balanced by having the resistors 41 and 48, 52 and 53, and 51 and-58 matched to produce in each stage plate and cathode voltages which are equal in magnitude and of opposite polarity. Likewise-the resistor, such asresistor 48, connected between the anodeand-operatingpotential in each stage, is much smaller than the resistor of the corresponding-phase shifting network, the resistor 50, for example. As an additional constructional detail of the network of Fig. 4, I have found that desirable operating characteristics are obtained when the time constants of the phase shifting networks of successive stages bear a ratio of approximately :1. Additionally, be cause of the unidirectional potential between a respective cathode and the control electrode of the succeeding tube, I have found that the oathode resistance of each stageshould have a value whichis equal to the number or order of the stage times the resistance of the cathode resistor of the first stage. Thus, resistor 58 in the The anode .of ,device 61 is,

terminals 10, II.

Range);

thirdstage has a value of resistance which is three times the value of resistor in the first I stage.

R7=R43=1000 ohms, balanced within 1%.

t2=- 5r=T2R .=2000ohms, balanced within 1%.

seconds. (R ='10 ohms.)

- R ta 413, 4000 ohms- R qc seeon s- (Depen s on audio mplifie l w freq e ts cu o -7 Dev ces 6. a d .5 are 55L? GT or S G ype ub s- For then network:

R47=R ;.=;10OO ohms, balanced within '1 seconds. (R 5 X10 ohmsi) Riv/F1 58 3.9.0.0 ohm b l d within 1%.

seconds. (R r-lfl ehms.)

1 togsoim secon s (Jim 5 19 ohms- Th .ztimeconstants of the product of resistor 43 and capacitor 44 in the .a and 3 networks are equal in order to insure .proper relative phase of input signals.

. Usin the va ues oithee mnonentseiven a ove, I.1h,av.e found t at the 5 network p ovid s a i .nal output app oximat lyfi .earlier in t m phase than oes the .a n two k when ach n w suppli d w th the sam signal a r ss ts pu e m nals re ations p, m re ver holds with a variat on of t rou h a freq ency range of B0 toYQQOcycles per second. While, in the circuit of Fig. 51, the signals available from the a. and 3 networks appear in push-pull form on conductors 64, 65,,resistor ,BZ may be shortireu ted o omitt d t provid a sin -e ded utput f om th respe tive cor I3 n two k.

T i cuit of. th esneotivemodulat s 24: emp oyed in the communic tion s stem of ,Fi 3 is shown in Big. '5 of thedrawings. This moduab ci u t com rises a pa r of penta rid converter electron discharge devicestfi, 61 having t ir cathodes connected t th r and s na of carrier wave frequency supplied in push-pull to the first control electrodes 68, 69 from input Th resisto s 1.2, 3 a d apacitors '14, 15 provide a coupling network to these control ,electrodes, while resistor 16, bypassed -by capacitor 1'! andhaving its ends connected respectively to the cathodes and the negative terminal of a battery 18 or any other suitable source of operating potential, serves to provide proper operating bias to the control electrodes 68, 69 of the device. Resistor "I9, having one terminal connected to the positive terminal of battery I8 and by-passed to the negative terminal by a capacitor 80, provides a suitable value of operating potential to the screen electrodes, the second and fourth electrodes of device 66, 61.

Push-pull modulating signals from the corresponding one of the a or 8 networks 2II23 are supplied to terminals 8|, 82 of the modulator circuit and to the respective third grids 83, 84 of devices 66, 61. I provide coupling networks comprising capacitor 85, resistance 86 and capacitor 3?, resistance 88 for the input terminals BI, 82 and choose the values of the impedance elements of these networks to translate all frequencies in the desired audio frequency band with negligible phase shift and amplitude discrimination. In this respect, I have found it desirable that the products of capacitor 85, resistor 86 and capacitor 81, resistor 88 be made equal.

The anodes 89, 90 of the respective devices 86, 81 are connected together and to a resonant circuit comprising an inductance 9I, a capacitance 92, and a resistance 93 connected in parallel. In this resonant circuit, which is tuned to the carrier frequency resistance 93 is of a value chosen to provide suincient band width to translate with negligible attenuation the sidebands corresponding to the highest frequency component of the modulating signals supplied to inputterminals BI, 82.

The circuit of Fig. 5 may represent any of the 1 modulation devices 242I of the system of Fig. 3. The anodes of the remaining modulator circuits may be connected to the tuned circuit comprising elements III-93 through conductor 94 and the combined output of the modulators appears on the terminal 95.

It is apparent from the modulator circuit of Fig. 5 that the carrier signals applied on conductors I0, II may be balanced so that no carrier appears on conductor 95. However, as explained previously in connection with the discussion of the circuit of Fig. 3, the amplifier 29 may be adjusted to supply any desired phase and magnitude of the carrier wave. Thus, the amplifier 29 may be a pentode amplifier whose anode is connected in common with the anodes f the modulators 242'I by means of the conductor 94 and whose control electrode is supplied with unmodulated carrier signal of desired phasefrom the oscillator I9. In this manner, the modulator circuit of Fig. provides in its output a signal which may be represented by Equation 33 when the proper carrier wave and modulating signals are supplied, respectively, to terminals 'IIL'II and8l,82.

Whila'in Fig. 5, I have illustrated one form of modulator circuit, it is obvious that other circuits may be employed for the modulators 24-2'I, provided they operate in accordance with the overall requirements of my communication system to supply to the power amplifier 39 an output wave which may be expressed by Equation 42. Thus, for example, I have found that acombination of four modulators containing but one pentagrid tube each may be made to'operate to meet'the requirements of my system, provided the correct signals are supplied to these modulators. Such a simplified group of modulatorahowever, is not inherently balanced and difficulty may be experienced in adjusting the group to satisfy Equa- '26 tion 42'over a wide band 'of modulating frequencies.

In Fig. 6, I have shown a major portion of the receiver circuit of the system of Fig. 3, together with a detailed circuit diagram of ,the exalted carrier detector 36 of the receiver of Fig.3. Fig. 6 gives certain circuit details of the arrangements shown in block form in Fig. 2 and elements of Fig. 6 corresponding, to elements of Fig; 2 are identified by corresponding reference numerals.

'In Fig. 6, the source of oscillations corresponding to the source A of Fig. 2 comprises a so-called electron coupled oscillator comprising an electron discharge device 96 and a frequency controlling or resonant circuit comprising an inductancev 91 and an adjustable'capacitance 98 having one terminal connected to the control electrode IOI of device 96 through a grid leak and condenser combination comprising resistor 99 and capacitor I09. The opposite terminal of the frequency controlling circuit 9'1, 98 is connected to ground and an intermediate point onthe inductance 9'1 is connected to the cathode ofdevice 96. Operating potential from the anode of the oscillator is provided from any suitable sourcev indicatedby the legend which is connected to the anode through a decoupling network comprising resistor I02 and capacitor I03. Device 96 isprovided with a stabilized screen voltage derived from a voltage divider comprising resistors I04 and I05, capacitor Iilt by-passing the screen electrode to ground at the operating frequency of the oscillator.

The phase shifter I2; comprises a tuned transformer having a primary winding I I11 and a secondary winding I 08, the primary winding being tuned by a capacitor H19 and the secondary by a capacitor III]. As iswell known, such a tuned transformer has the property of producing a 90 phase shift between theprimary and secondary voltages at itscenter frequency when both windings are properly tuned. Accordingly, by controlling the coupling between the windings of this transformer, the oscillations or carrier voltages from the local oscillator 96 appearing at the output terminal of the secondary winding I238 may be of the same magnitude asthat of the input terminal of the primary winding I01, but displaced in phase therefrom by 90. I provide loadin resistors III, I I2 connected respectively across the primary and secondary windings to improve the performance of the exalted carrier detector circuit by extending the range over which the oscillator frequencymay be allowed to vary. As a result, when the amount of loading furnished by the resistors III, H2 is properly adjusted,

slight mistuning of the receiver has less effect on the 90 phase shift-obtained in the transformer. In general, when the values of resistorsl I I, II2 are small, the isolation or decouplingbetween the frequency controlling oscillator circuit 9?,1981and the phase shifting device, I2 is increased and the amount of signal voltage available at theter= minals of the windings I 0'1, l l 8 is reduced;

. The diodes I, 2 are illustrated as beingincluded in a single. envelope and are coupled through capacitor 3 to the input terminal of primary winding IB'Ito improve thestability'of the servo loop circuit controllinegthephase of the signal applied to diodes I and 2: from thesource A relative to the phase of the incoming signal from source B. j

The tuning device I3 comprises an electron discharge device b having connectedbetween its anode and cathode-a phase shiftin circuit comprising capacitor H4 and resistor H5; the

l 'vide' si'gnal from theTcircuit 91598 to the phase 'shiftingcircuit i I I; I I 5. 4 "The signal applied from I phase 'shifting circuit i I 4, I I5to the f control electro'd or device H3: is fapproirimately 90-out of i PhasewithI-thatatthe anodeof device H3 and i causes anodeicurrentitd i'fiow in'su'ch a manner -.a's t'o"make-theimpedancepresented by thereactance -device I3 to the"frequency controlling circuit "91, -'98 appear reactive. The amount of such reactance 'present'edto the frequency controlling circuit of-the source A by': the rea'ctance'"d'evice I3 iSQQOh-trolled'bj'th 'blas 'potentialsupplied t the c'ontrol electrode ofdevicei I3 over conductor ls c'onneotedzto the-commonpoint or r'esistor'l 4 and capacitor 5. Operating potential -''for the anodebi devlce I I 3' is -supplied through plate resistor H 1- rrom the sourc'of 'p'otntial indicated b'y'the'legend-+.

: A st'abilized source-"of bias' potential is supplied -to the cathode of device' I I3-from av'oltag divider circuit:comprising'iresistors 7 H 8, I I9 connected in series 'betweenthe" sourceof -'potential and ground,'the"common point otthese resistors being connected to thejcathode of device-I I3 ahdcapacitor I20, serving as aradi frequency bypass; be-

I ing' connected across resistance H9. Asimilar stabilizedis'ource orpctentiar-is provided for the 'screen electrode of device I I3 by-voltage dividing resistors-'- I 2 I; I2 2 connected in' series between the source 'and' groundj the screen electrode being 'connected to the "common point of these resistjan'cesandbeing'by passed to'ground' by capacitor I23. Alternatively, a-iconv'entionalvoltage-regulator tube of th gaseousdischarge type may be employed to stabilize the potential of the screen electrodes "of device's l l3 and 96.

The portion 'of 'the-circuitof 'Figi 6-Which is labelled 's'ou'rceB and-corresponds to the rec- "tangle bearing the numeralf inf 'Fig'. 2 com- I prises "receiving antenna" 32,-"-'radio receiving-amplifler and mixer stage 33, the-local-oscillator-34,

' and the" F interniedi'atfrequency amplifier- 35.

Theo'utput of "the intermediate "frequency amplifier' is coupled to an intermediate frequency transformer comprising a primary winding "I24 and a secondary'v/inding I25=thesewindings-be- 5 ing tuned respectively to 'the intermediate'frequency wi by capa'citors I26, I'ZII'On'e terminal cit-secondary winding lzi isconnectcd to'thecontrol electrode'of an electron discharge-device I 2 8 .and' the "opposite terminal" is supplied with a source orbi'as by-connection'tothe commenterminal or Voltagedividing' resistances I 2 9, "I the "resistor I29' being shuntedby a" capacitor "I3I. Operating potential for-device "128- is provided through a decouplingnetw'ork comprising resistor I32 connected to a source of positive" potential "and by-pass capacitor I33 connected to ground. Device -I28op erates as a cathode follower having a'resi'stance I34, connected-between the'cathode and ground, to provide" alow'impedance source ofsignals otthefrequen'cy & 2x

to the cathodes or the diodes I, I'.

As explained'in thedl'scu'ssionof the circuit of Fig. 2, the'combination of resistorI4,Icapacitor'I 5 functions *as; a1.filterwh'ichsipermitsxonly.:slow

variations f'oft: potential 'ifiomcconductorrl 6. :The operation of thi i filterneircuit: is stabilized iby i meanslof atnetwcrk'.sliuntediacross capacitoi'cl 5 and compnisinga res'istorlflkzand capacitor m.

Simi1a3r1y; the networks rcomprisingnresistances I 4', l 3I and capacitors'l 5" ,n I:3B="serve:\to.-provide a stabilizedlfiltered 'unidirectionarzpotentialapropo'rtional to thescarrier'component of the-signal supplied by device I2 8 to"the scathodesmt diodes I I 1 To fthis e'nd,':the L's-unidirectional Ipotential developed I acrossrcapacitor I 5'nm'ayzbe supplied to Ithe" radiofirequency. ampiifiercand zmixer 33 and intermediate: frequency amplifienuwi automatically to. controlithetg'ain :of thesetcircuitsdn a conventional: rnannerxslntthis network iresistor I31- andfcapacitorzl33feonnectedzacnoss capacitor I 5' provi'de a 'loadi'circuitrfor diodes" .I is," 2 'which is identical to thatifemployedzinathg :eircuit;:of

diodes I, 2. The output=currentsiofthe exalted carrierdetector'are supplied byicon'ductors II, II t'othe a and fl-netWorksI'lI, 3iiasexplained'inzthe discussion of the'system oi'fr'ig'. 3.

' One circuit foi 'ithe' differentialmmplifierr33 of my improvedre'ceiver illustratedfin'Fig'. 7 and comprises a pair of electron discharge-:devices I-H, I40 having their cathodes connected to ground through a common mesistance MI and their anodes connected, respectively, to a source of unidirectional potential through "equal resistances 'I42','-I43. The outputterminal;:of .the ametwork 31 is coupled to the control electrode ofhdevice I 39 through a capacitorI44 and theoutputterminal of the ,8 network -'38 is coupled-to the -control electrode of -the'device 440 through a-capacitor =I'45. The-control electrodesare likewise connected, V respectively, through i equal --resistances I46, I 41 to the commonterminal ot-voltage dividing resistances I 48-,- I49,- -the latter resistance being by-passed to-ground vby acapaciton I5i'l. .lThe resistances I48, I49 are connected across .the

' Source of operating potential totsupply-ldesired bias potential to: the control electrodes: of -the devices 'I 39,-1'40.

In the-operation'of the differential-amplifier in a conventional-manner,theoutput terminal l 5 I coupled to-the cathodes of devices I33-andsI'40,

. provides a signal which comprises-the sum: of the signals supplied 'by the a and B=networks. =Ihe 'equation-of'the sumof these-"two signalsis that t given by Equation 50and 'consistsisolely--of'-the intelligence 1). On-the otherrhand; the -si'gnal-on terminal-I 52 coupled 'to'. the anode -orrdevice l'4fl comprises the difierence' betweenwthe signals-of theoutputs of -the sand 5 networkslil'Ihiswdifference signaLthz-expression of: which=is glven I in Equation 5lpc0nsists solely ofintelligence q.

1 In :Fig; 8, I-:have1shown:amaltemative vcix'cuit for'the difierehtiatamplifieriB 9? which: comprises a simple resistance mixing circuitxl'lr'ilFig'. 8,'the

' output:terminals-lt4l for .the ti networl'r' fl are of potentiometers': I53,-I 54; The -rcspective teri minals 64 -65 'are connected with 'therpotentiom- T eters I53, T I 54 through c'ou'pling capacItors ISS to keep the unidirectional currents in 'the 'outputs of the it and e -networks-rroni' flowing" through 'the'potentiometers.

I The-terminal 5550f the B-netw0rk -38'is coupled tothe other terminals "or-'- these potentiometers. Ifthe values ofrsistor Stand 63 in the a and [3 networks are thosegiven previously' in a table of values for the elements of the circuit 'o'f Figj 4,

'the signal potentials-on cOnductQrS' MJ- GEare balanced to'- ground: tor b'oth netwcrks. Accord- .ingly, by, the connectionsshown in Fig. ..8,.a signal equal to the'sum of thesign'als in the .a, and 6 networks, and which may, comprise intelligence p only, may be obtained from tap [56, on potentiometer 154. A signal equal to. differenceof the signals in the outputs of the. ,aand B networks, and vwhich may comprise signals. of q intelligence only, is available onthe tap: l5liof potentiometer I53. These taps l56,:l15'l are. movable so that they provideaj means ioribala'ncing or .adjusting the signals, supplied, respectively, to; the amplifiers 40, Al. mp-i;

; In the system shown-in block.diagramin Fig. 3,

. the si nals. at th output orthe pairs, of ands networks oithe, transmitterha'veconstant amplitud s an While their phase jdiffierenc je-imay vary sl htly from .9Qfl.due... o. theifactuthatth Island. 3 n tworksam .not...perf.ect;gnetworks, I h ve f und-that :py. suitably p portion ne t values of the components of these networksthis; deviation from 90 phase relation may bemade ve y s the ma mum deviation being or. the order of 2. or lessin translating a ,band ofirequencies the maximum frequency of which is a pro a 1y..l.0 mes-.: t e mi imum. irequency. In the system of Fig. 9, I have shown a modification of; selective; sideband system n w h e pa rzo voltaeesu d. on d ins a i wa eiare mai .p n d:- x c ya a ons n 9 phase. d filerenceb t flw i h a Sli ht va at on-1m: ma nitud f-these volta may be encountered because. of the imperfect a ndfi. n tw rks 'e o m'rhe. sys em (ifi 9 is substantially the same as; the system of Fig- 3 and elementsthereof corresponding to elements of the system of Fig. 3 are;indicated by correspendingreferencenumerals... V I a In the transmitter portion of the system of Fig. ,9, a difierentialamplifier I58 is connected between the w and 5. networks 20,; 2! and the modulators 2,4, 25,v Another differential amplifier I59 is connected between; the e ands networks 22, 23. and modulators 26, 21;. The-remaining elements of the transmitter are identical-with th s us d th syst m of F 3- the r ceiver, elements 3 3, 34 and3 are similar to that shown in Fig. 6, Whereintheoutput-of- I. F. am-

plifier 35 comprises odulated carrier waves at the'new oarrier frequency s produced" as "a" result" of heterodyning the oscillations of 34 with the incoming waves available over" antenna" 321 and including the upper and lowerf'side bands ii bearing th intelligence of channels 10 and q. Thedetector' 36 whi'ch is shown in- Fig. 6 imixesithe 'output of 'a' local source of oscillation 'shown in block 4 of Fi .6 with the upper and lower side bands transmitted from the immediate frequency of"the' amplifier 35 before application to the diiferential amplifier I69 shown in Fig. 9. The differential amplifier "I653 is connected between the combined exalted carrier detectors "36 and the sand 5 networks 31,38. 1 While .the operation of the system of Fig. 9 may be explained by rigorous mathematical analysis similar to thatemployed in discussing the operation of theJsystemof "Fig.8, I have chosen toexplain the'operation of the'system of Fig. 9ibyzthevector diagramsshown in Figs. 10-15.. :Of, course a similar 'vectorial analysis could be applied to thexsyst'em'of Fig. 3.

.. may represent the voltageat; theeoutput... 02. the

' at the 'modulation frequency a network 20 and the vectorlbearingj the .legend M may represent the voltage at the output of the 18 network 2]. Due to the imperfect characteristics inherent in such a and 5 networks, there two voltages, instead of being displacedin phase by exactly may vary from -theQO phase relationship by the angle 6, but will have-exactly equal amplitudes- In the differential amplifier, the vector voltage Ma, Mp are added to produce the voltage 'represented-byithe;-vecto1 bearing the legend M cospt." Thevectorwoltage-M is subtracted from thevoltag in this sameamplifier to produce the vector voltage bearingthe legend 1 sin ptWI'he voltageM cos pt is displaced from the voltage Mil-sin pt by exactly 90. However, the. scalar magnitudes M', M" are not equal but vary by a sm all amount depending on the characteristics' of the a and-13 networks through which they have passed.

The components of the voltage at the output of modulator 24 are shown in the vector diagram of Fig. 11. These components-comprise thecarrier Wave cos wt, the"lower.sideband-signab 1 cos .andthe upper sideband coiripohent.r

V 2" v If the modulator 274 comprises; the ,fcircuit, of Fig. 5, the carrier component cos wt does not appear in the output but is suppressed due to the operation of the modulator circuit." However, the carrier componentis shown in Figs. ll-15 to e'stabli'sha reference base Iorpurposs-of easy visualization." Moreoverfi Figs. 11 l5 the vector representing the carri l lcomponent lisfconsiderd as'staaonar and insist-aris :of the other"vectois' relative to the earner component "only are considered. Ofco'ur efjhe rotation oi' the vectors representing the"modulationjcomponents relative to thefcarrier oniponent occurs [In the-mod lator amine, titer-tan ismodulatedbythe vector 'vonage'M" s13 tfto produce and output voltage, the vector diagram or which is showni'n Fig". 12; This voltage comprises a carrier component "sing at plus] two equ l and oppositely rotating modulation" campai ns cqillpiising e l litPaP .PQhFP Q QtQ 2 and the up er sideband component g cos (w+p)t to produce a desired resulta-ntsideband MI/MI Q z si pi l:

rile: upper sidebaiid co nponents er er pnase and, when aadedlttgthmnreefiee -firie vector voltage, illustrated in Fig.- 141 by the small vector bearingthe legend I I I/ This resultant voltage is an undesired voltage and arises because of the unequal magnitudes of the vector volt-ages M, M. The difference AM between these two volt-ages is given by the equation M 'M"=AM 'Ihe radiated signalincluding a carrier from the amplifier 29 may be illustrated by. the .vector shown in the left-hand portion of Fig. 15 and comprises the carrier component cos wt, the de sired lower sideband component I I/ 1 WW re) and the undesired-component cos (w+p)t This vector is identically equalto that shown in the right-hand portion of Fig. in which the component voltages are .given in terms of vectors M" and AM. Thus, the volt-age as illustrated in the right-hand portion of Fig. 15, may be represented as comprising the carrier component cos wt. the desired lower sideband component M" cos (w29)t, and the undesired,

component made up of two oppositely rotating vectors having values equal, respectively, to

signals fromthe channel q likewise are considered, similar analysis shows that the radiated signal contains components M cos (w+q)t and undesired components Assuming that the upper sideband is used as the principal sideband for the transl-ation'o'fthe q intelligence, it can be shown'th'at the output of the one of the detectors 36 in the receiver, which for the p intelligence provided thesignal M cos pt, gives for the q intelligence the signal (M +AMq) cos qt=M cos qt. The output of the other detector of the combined exalted carrier detectors 36 is Mq" sin qt.

For either channel 1) or q, the differential amplifier I60 in the receiver converts the M and M" signals to M3 and a signals so that the a and 5 networks 31, 38 can provide ideal separation of the 9 intelligence from the q intelligence and 'supp1y-.,the proper intelligence:to

the :channel amplifiers 40, 4|. It is obvious, of

course, that, in the receiver of both Figs. 3 and 9, any desired switching arrangement may be employed for selecting either or both oLthe upper and lower sidebands to receive the intelligences translated by the selected sideband. Thus, such switching arrangements may beincluded in the. differential amplifier 39 or between that amplifier and the channels 40, 41. Alternatively, both intelligences maybe-received simultaneously to produce binaural effects where the sources at the transmitter were exposed to the same sound field. Likewise, similar switching arrangements may be employed in the transmitters of the sources of Figs. 3

and 9 so that both sidebands may be transmitted simultaneously or any desired sideband may be transmitted alone, i. e., either the upper sideband or the lower sidebandmay be-used by itself to transmit the desired intelligence signals.

An important advantage of the improved communication systems described above is that they permit simultaneous single sideband modulation of a radio frequency carrier by two sources of intelligence and reception of the resulting composite signal with apparatus which sepa rates the two intelligence channels and provides an output of each on different pairs of terminals. Thus, the overall system may be used for multiplex transmission and reception of any two channels which may be any type of communication or entertainment without requiring, ii the bandwidths of the two channels are the same, a greater bandwidth than that required for either channel alone whenconventional double sideband transmission and reception are employed.

In the transmitters of the systems disclosed in Figs. 3 and 9, binaural signals may be radiated,

which signals may be received by conventional receivers now in use and translated by these receivers in their normalfashion. On the other hand, these signals may be received by a binaural receiver to produce binaural effects in the output thereof. The binaural receivers described above, morover,-employ noexpensive filters, so that they are relatively simple in operation-and inexpensive in construction. It is thus apparent that by my invention it is possible to have a radio broadcasting arrangement in which both normal single channel transmitters and double.

single sideband transmitters may be operated. Either/conventional receivers or my improved receiver-circuits may be employed to produce,

respectively, conventional reception effects or binaural effects. Thus,v the systems disclosed permit a transition from the present broadcasting system, to a new and improved arrangement without, obsoletingexisting equipment in the sense that such existingequipmentis made useless during such transition.

It isapparentlikewise that, in my improved system, since no filters are employed, the circuit arrangements are not limited by considerations normally encountered when filters are used; that is, the transmitters may utilize both very low andwery high modulation frequencies and the receivers produce faithful reproduction of all such modulation frequencies.

-In my improved transmitters, when the two intelligence channels are made identical, it is apparentthat the'two sidebands'may be made to bear the same phase :relations relative to the carrier as; :those: produced? by a. conventional 

